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MAX16975 28v, 1.2a automotive step-down converter with low operating current ????????????????????????????????????????????????????????????????? maxim integrated products 1 typical application circuit 19-5673; rev 0; 3/11 ordering information appears at end of data sheet. general description the MAX16975 is a 1.2a current-mode step-down con - verter with an integrated high-side switch. the device operates with input voltages from 3.5v to 28v while using only 45 f a quiescent current at no load. the switching frequency is adjustable from 220khz to 1.0mhz by using an external resistor, and can be synchronized to an exter - nal clock. the devices output voltage is pin-selectable to a fixed 5v or adjustable from 1v to 10v using external resistors. the wide input voltage range makes the device ideal for automotive and industrial applications. the device operates in skip mode for reduced current consumption in light-load conditions. an adjustable reset threshold helps keep microcontrollers alive down to the lowest specified input voltage. protection features include cycle-by-cycle current limit, soft-start, overvolt - age, and thermal shutdown with automatic recovery. the device also features a power-good monitor to ease power-supply sequencing. the device operates over the -40 c to +125 c automo - tive temperature range, and is available in 16-pin qsop and qsop-ep packages. features s wide 3.5v to 28v input voltage range s 42v input transient tolerance s 5v fixed or 1v to 10v adjustable output voltage s integrated 1.2a high-side switch s 220khz to 1.0mhz adjustable switching frequency s frequency synchronization input s internal boost diode s 45a skip-mode operating current s less than 10a shutdown current s adjustable power-good output level and timing s 3.3v logic level to 42v compatible enable input s current-limit, thermal shutdown, and overvoltage protection s -40c to +125c automotive temperature range applications automotive industrial for related parts and recommended products to use with this part, refer to: www.maxim-ic.com/MAX16975.related d1 r fb1 25ki r fb2 100ki c out1 47f c out2 47f c in2 4.7f c cres 1nf r comp 12ki r res 10ki r fosc 61.9ki l1 10h v out = 1.25v at 1.2a at 400khz c bst 0.1f lx bst v out out 3.5v to 28v fb reseti v bias res fosc cres c bias 1f c comp2 open bias c comp1 5600pf comp fsync en supsw sup gnd c in1 47f c in3 0.1f MAX16975 place c in3 (0.1f) right next to sup. for pricing, delivery, and ordering information, please contact maxim direct at 1-888-629-4642, or visit maxims website at www.maxim-ic.com.
????????????????????????????????????????????????????????????????? maxim integrated products 2 MAX16975 28v, 1.2a automotive step-down converter with low operating current stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. these are stress ratings only, and functional opera - tion of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. exposure to absolute maximum rating conditions for extended periods may affect device reliability. electrical characteristics (v sup = v supsw = 14v, v en = 14v, l1 = 22 f h, c in = 4.7 f f, c out = 100 f f, c bias = 1 f f, c bst = 0.1 f f, c cres = 1nf, r fosc = 61.9k i , t a = t j = -40 n c to +125 n c, unless otherwise noted. typical values are at t a = +25 n c.) sup, supsw, lx, en to gnd ............................... -0.3v to +45v bst to gnd .......................................................... -0.3v to +47v bst to lx ............................................................... -0.3v to +6v out to gnd .......................................................... -0.3v to +12v sup to supsw ..................................................... -0.3v to +0.3v reseti, fosc, comp, bias, fsync, cres, res , fb to gnd ......................... -0.3v to +6v output short-circuit duration .................................... continuous continuous power dissipation (t a = +70 n c) qsop (derate 9.6 mw/ n c above +70 n c) ................. 771.5mw operating temperature range ........................ -40 n c to +125 n c junction temperature ..................................................... +150 n c storage temperature range ............................ -65 n c to +150 n c lead temperature (soldering, 10s) ................................ +300 n c soldering temperature (reflow) ...................................... +260 n c absolute maximum ratings note 1: package thermal resistances were obtained using the method described in jedec specification jesd51-7, using a four- layer board. for detailed information on package thermal considerations, refer to www.maxim-ic.com/thermal-tutorial . qsop junction-to-ambient thermal resistance ( q ja ) ..... 103.7c/w package thermal characteristics ( note 1) parameter symbol conditions min typ max units supply voltage v sup , v supsw normal operation 3.5 28 v supply current i sup normal operation, no switching 2.9 ma skip mode, no load, v out = 5v 45 f a shutdown supply current v en = 0v 9 f a bias regulator voltage v bias v sup = v supsw = 6v to 42v, v out < 3v or v out > 5.5v, i load = 0a (note 2) 4.7 5.0 5.3 v bias undervoltage lockout v uvbias v bias rising 2.95 3.15 3.35 v bias undervoltage hysteresis 550 mv thermal-shutdown threshold +175 n c thermal-shutdown threshold hysteresis +15 n c output voltage (out) output voltage v out normal operation, v fb = v bias , i load = 1a, t a = +25 c 4.95 5 5.05 v normal operation, v fb = v bias , i load = 1a, -40 c p t a p +125 c 4.9 5 5.1 skip-mode output voltage v out_skip no load, v fb = v bias (note 3) 4.9 5.05 5.2 v load regulation v out = 5v, v fb = v bias, 30ma < i load < 1a 0.3 % line regulation 6v < v sup < 28v 0.02 %/v * the parametric values (min, typ, max limits) shown in the electrical characteristics table supersede values quoted elsewhere in this data sheet.
????????????????????????????????????????????????????????????????? maxim integrated products 3 MAX16975 28v, 1.2a automotive step-down converter with low operating current electrical characteristics* (continued) (v sup = v supsw = 14v, v en = 14v, l1 = 22 f h, c in = 4.7 f f, c out = 100 f f, c bias = 1 f f, c bst = 0.1 f f, c cres = 1nf, r fosc = 61.9k i , t a = t j = -40 n c to +125 n c, unless otherwise noted. typical values are at t a = +25 n c.) parameter symbol conditions min typ max units bst input current i bst v bst - v lx = 5v 1.7 2.5 ma lx current limit i lx v sup = 4.5v to 28v, v supsw = 14v, t a = +25 c 1.5 1.8 2.0 a v sup = 4.5v to 28v, v supsw = 14v 1.5 1.8 skip-mode current threshold i skip_th 200 ma power-switch on-resistance r on r on measured between supsw and lx, i lx = 1a, v sup = 4.5v to 28v, v bst - v lx = 4.5v 300 550 m i lx leakage current i lx,leak v supsw = 28v, v lx = 0v, t a = +25 c 0.01 1 f a transconductance amplifier (comp) fb input current i fb 20 na fb regulation voltage v fb fb connected to an external resistive divider, t a = +25 c 0.99 1.0 1.01 v fb connected to an external resistive divider, -40 c p t a p +125 c 0.985 1.0 1.015 fb line regulation d v line 4.5v < v sup < 28v 0.02 %/v transconductance (from fb to comp) g m v fb = 1v, v bias = 5v 1000 f s minimum on-time t on 110 ns cold-crank event duty cycle dc cc 94 % oscillator frequency oscillator frequency r fosc = 25.5k i , v sup = 4.5v to 28v 1.0 mhz r fosc = 61.9k i , v sup = 4.5v to 28v 348 400 452 khz r fosc = 120k i , v sup = 4.5v to 28v (note 3) 191 220 249 khz oscillator frequency range f osc (note 3) 220 1000 khz external clock input (fsync) external input clock acquisition time t fsync 1 cycles external input clock frequency (note 3) f osc + 10% hz external input clock high threshold v fsync_hi v fsync rising 1.4 v external input clock low threshold v fsync_lo v fsync falling 0.4 v fsync pulldown resistance r fsync 500 k i soft-start time t ss f sw = 400khz 4 ms f sw = 1.0mhz 1.6 ms enable input (en) enable on threshold voltage low v en_lo 0.8 v * the parametric values (min, typ, max limits) shown in the electrical characteristics table supersede values quoted elsewhere in this data sheet.
????????????????????????????????????????????????????????????????? maxim integrated products 4 MAX16975 28v, 1.2a automotive step-down converter with low operating current electrical characteristics* (continued) (v sup = v supsw = 14v, v en = 14v, l1 = 22 f h, c in = 4.7 f f, c out = 100 f f, c bias = 1 f f, c bst = 0.1 f f, c cres = 1nf, r fosc = 61.9k i , t a = t j = -40 n c to +125 n c, unless otherwise noted. typical values are at t a = +25 n c.) note 2: when 3v < v out < 5.5v, the bias regulator is connected to the output to save quiescent current, v bias = v out . note 3: guaranteed by design; not production tested. parameter symbol conditions min typ max units enable on threshold voltage high v en_hi 2.2 v enable threshold voltage hysteresis v en,hys 0.2 v enable input current i en 10 na reset reset internal switching level v th_rising v fb rising, v reseti = 0v 93 95 96.5 %v fb v th_falling v fb falling, v reseti = 0v 91 93 95 reseti threshold voltage v reseti_hi v reseti falling 1.05 1.25 1.4 v cres threshold voltage v cres_hi v cres rising 1.07 1.13 1.19 v cres threshold hysteresis v cres_hys 0.05 v reseti input current i reset v reseti = 0v 0.02 f a cres source current i cres v out in regulation 9.5 10 10.5 f a cres pulldown current i cres_pd v out out of regulation 1 ma res output low voltage i sink = 5ma 0.4 v res leakage current (open- drain output) v out in regulation t a = +25 c 1 f a t a = +125 c 20 na reset debounce time t res_deb v reseti falling 25 f s * the parametric values (min, typ, max limits) shown in the electrical characteristics table supersede values quoted elsewhere in this data sheet.
????????????????????????????????????????????????????????????????? maxim integrated products 5 MAX16975 28v, 1.2a automotive step-down converter with low operating current typical operating characteristics (v sup = v supsw = 14v, v en = 14v, l1 = 4.7 f h, c in = 4.7 f f, c out = 22 f f, c bias = 1 f f, c bst = 0.1 f f, c cres = 1nf, r fosc = 61.9k i , t a = +25 n c, unless otherwise noted.) startup with full load (out = 1.25v, f sw = 400khz) MAX16975 toc01 0v 0v 0v 0v out 1v/div res 5v/div en 5v/div sup 5v/div 2ms/div i load = 1.2a efficiency vs. load current MAX16975 toc02 i load (ma) 400 800 1200 10 20 30 40 50 60 70 80 90 100 0 0 8v/400khz 5v/400khz 3.3v/400khz 1.25v/400khz efficiency (%) switching frequency vs. load current (1.25v/400khz) MAX16975 toc03 i load (ma) switching frequency (khz) 950 700 450 400.4 400.8 401.2 401.6 402.0 400.0 200 1200 pwm mode switching frequency vs. r fosc MAX16975 toc04 r fosc (ki) switching frequency (khz) 98 76 54 32 400 600 800 1000 1200 200 10 120 5v output switching frequency vs. temperature (1.25v/400khz, 5v/400khz) MAX16975 toc05 switching frequency (khz) 370 390 410 430 450 350 temperature (c) 110 95 80 65 50 35 20 5 -10 -25 -40 125 i load = 1.2a, r fosc = 64.87ki 1.25v/400khz 5v/400khz load-step response (1.25v/400khz) MAX16975 toc06 0 0 v out 100mv/div i load 1a /div 4ms/div 0 to 1.25a load step v out ac-coupled
????????????????????????????????????????????????????????????????? maxim integrated products 6 MAX16975 28v, 1.2a automotive step-down converter with low operating current typical operating characteristics (continued) (v sup = v supsw = 14v, v en = 14v, l1 = 4.7 f h, c in = 4.7 f f, c out = 22 f f, c bias = 1 f f, c bst = 0.1 f f, c cres = 1nf, r fosc = 61.9k i , t a = +25 n c, unless otherwise noted.) cold-crank pulse (1.25v/400khz) MAX16975 toc07 0v 0v 0v 0v v supsw 10v/div v out 1v/div v res 5v/div v lx 10v/div 10ms/div dips and drops test (1.25v/400khz) MAX16975 toc08 0v 0v 0v 0v v supsw 10v/div v out 1v/div v res 5v/div v lx 10v/div 10ms/div slow v in ramp-up test MAX16975 toc09 v sup/ supsw 10v/div v out 5v/div v lx 10v/div 10s/div output short-circuit test (1.25v/400khz) MAX16975 toc10 0v 0v 0a v out 2v/div i load 2a/div v lx 10v/div 1ms /div r load = 0.3i quiescent current vs. input voltage MAX16975 toc11 input voltage (v) quiescent current (a) 24 20 16 12 8 10 20 30 40 50 60 70 80 90 0 4 28 5v/400khz v out vs. temperature in pwm mode (5v/400khz) MAX16975 toc12 output voltage change (%) -1 0 1 2 -2 temperature (c) 110 95 80 65 50 35 20 5 -10 -25 -40 125 i load = 1.2a 5v/400khz
????????????????????????????????????????????????????????????????? maxim integrated products 7 MAX16975 28v, 1.2a automotive step-down converter with low operating current typical operating characteristics (continued) (v sup = v supsw = 14v, v en = 14v, l1 = 4.7 f h, c in = 4.7 f f, c out = 22 f f, c bias = 1 f f, c bst = 0.1 f f, c cres = 1nf, r fosc = 61.9k i , t a = +25 n c, unless otherwise noted.) v out vs. temperature in pwm mode (1.25v/400khz) MAX16975 toc13 output voltage change (%) -1 0 1 2 -2 temperature (c) 110 95 80 65 50 35 20 5 -10 -25 -40 125 i load = 1.2a v out vs. temperature in skip mode (5v/400khz) MAX16975 toc14 output voltage change (%) -1 0 1 2 -2 temperature (c) 110 95 80 65 50 35 20 5 -10 -25 -40 125 i load = 0a, skip mode 5v/400khz v out vs. temperature in skip mode (1.25v/400khz) MAX16975 toc15 output voltage change (%) -1 0 1 2 -2 temperature (c) 110 95 80 65 50 35 20 5 -10 -25 -40 125 i load = 0a, skip mode line regulation MAX16975 toc16 v out (v) 4.95 5.00 5.05 5.10 4.90 v supsw (v) 26 24 22 20 18 16 14 12 10 8 6 28 5v/400khz
????????????????????????????????????????????????????????????????? maxim integrated products 8 MAX16975 28v, 1.2a automotive step-down converter with low operating current pin configurations pin description pin name function 1 cres analog reset timer. cres sources 10 f a (typ) of current into an external capacitor to set the reset timeout period. reset timeout period is defined as the time between the start of output regulation and res switch - ing to high impedance. leave cres unconnected for minimum delay time. 2 fosc resistor-programmable switching frequency control input. connect a resistor from fosc to gnd to set the switching frequency (see the internal oscillator section). 3 fsync synchronization input. the device synchronizes to an external signal applied to fsync. the external signal period must be 10% shorter than the internal clock period for proper operation. 4 i.c. internally connected. connect to gnd. 5 comp error-amplifier output. connect a compensation network from comp to gnd for stable operation. see the compensation network section. 6 fb feedback input. connect an external resistive divider from fb to out and gnd to set the output voltage between 1v and 10v. connect fb directly to bias to set the output voltage to 5v. see the applications information section. 7 out connect out to the output of the converter. out provides power to the internal circuitry when the output voltage of the converter is set between 3v and 5.6v. during shutdown, out is pulled to gnd with a 50 i resistor. 8 gnd ground 9 bias linear regulator output. bias powers the internal circuitry. bypass bias with a 1 f f capacitor to ground as close as possible to the device. during shutdown, bias is actively discharged through a 32k i resistor. 10 bst high-side driver supply. connect a 0.1 f f capacitor between lx and bst for proper operation. 11 sup voltage supply input. sup powers the internal linear regulator. connect a 4.7 f f capacitor from sup to ground. connect sup to supsw. 12 lx inductor connection. connect a rectifying schottky diode between lx and gnd. connect an inductor from lx to the output. en supsw i.c. 1 2 16 15 reseti res fosc fsync cres top view 3 4 14 13 sup bst out 5 1 2 lx comp fb 6 7 11 10 bias gnd 8 9 MAX16975a + qsop en supsw i.c. 1 2 16 15 reseti res fosc fsync cres top view 3 4 14 13 sup bst out 5 1 2 lx comp fb 6 7 11 10 bias gnd 8 9 MAX16975b + qsop ep
????????????????????????????????????????????????????????????????? maxim integrated products 9 MAX16975 28v, 1.2a automotive step-down converter with low operating current pin description (continued) functional diagram pin name function 13 supsw internal high-side switch supply input. supsw provides power to the internal switch. connect a 4.7 f f capacitor from supsw to ground. connect sup to supsw. see the input capacitor section. 14 en battery-compatible enable input. drive en low to disable the device. drive en high to enable the device. 15 res open-drain active-low reset output. res asserts when v out is below the reset threshold set by reseti. 16 reseti reset threshold level input. connect to a resistive divider to set the reset threshold for res . connect reseti to gnd to enable the internal reset threshold. ep exposed pad (MAX16975a only). connect ep to a large-area contiguous copper ground plane for effec - tive power dissipation. do not use as the only ic ground connection. ep must be connected to gnd. 10a comp comp b.g. ref soft- start uvlo ldo standby supply ref ea logic for 100% duty-cycle operation res v bias fb reseti cres pwm comp level shift i lim logic en osc sum i sense fsync fosc bias sup out comp drv lx bst supsw gnd MAX16975 mux
???????????????????????????????????????????????????????????????? maxim integrated products 10 MAX16975 28v, 1.2a automotive step-down converter with low operating current detailed description the MAX16975 is a constant-frequency, current-mode automotive buck converter with an integrated high-side switch. the device operates with input voltages from 3.5v to 28v and tolerates input transients up to 42v. during undervoltage events, such as cold-crank condi - tions, the internal pass device maintains 94% duty cycle for a short time. an open-drain, active-low reset output helps to monitor the output voltage. the device offers an adjustable reset threshold that helps to keep microcontrollers alive down to the lowest specified input voltage and a capacitor- programmable reset timeout to ensure proper startup. the switching frequency is resistor-programmable from 220khz to 1.0mhz to allow optimization for efficiency, noise, and board space. a clock input, fsync, allows the device to synchronize to an external clock. during light-load conditions, the device enters skip mode that reduces the quiescent current down to 45 f a and increases light-load efficiency. the 5v fixed output voltage eliminates the need for external resistors and reduces the supply current by up to 50 f a. linear regulator output (bias) the device includes a 5v linear regulator, v bias , that provides power to the internal circuitry. connect a 1 f f ceramic capacitor from bias to gnd. when the output voltage is set between 3v and 5.5v, the internal linear regulator only provides power until the output is in regula - tion. the internal linear regulator turns off once the output is in regulation and allows out to provide power to the device. the internal regulator turns back on once the external load on the output of the device is higher than 100ma. in addition, the linear regulator turns on anytime the output voltage is outside the 3v to 5.5v range. external clock input (fsync) the device synchronizes to an external clock signal applied at fsync. the signal at fsync must have a fre - quency of 10% higher than the internal clock frequency for proper synchronization. adjustable reset level the device features a programmable reset threshold using a resistive divider between out, reseti, and gnd. connect reseti to gnd for the internal threshold. res asserts low when the output voltage falls to 93% of the programmed level. res deasserts when the output voltage goes above 95% of the set voltage. some microprocessors accept a wide input voltage range (3.3v to 5v, for example) and can operate during dropout of the device. use a resistive divider at reseti to adjust the reset activation level ( res goes low) to lower levels. the reference voltage at reseti is 1.25v (typ). the device also offers a capacitor-programmable reset timeout period. connect a capacitor from cres to gnd to adjust the reset timeout period. when the output volt - age goes out of regulation, res asserts low and the reset timing capacitor discharges with a 1ma pulldown current. once the output is back in regulation the reset timing capacitor recharges with 10 f a (typ) current. res stays low until the voltage at cres reaches 1.13v (typ). dropout operation the device features an effective maximum duty cycle to help refresh the bst capacitor when continuously oper - ated in dropout. when the high-side switch is on for three consecutive clock cycles, the device forces the high-side switch off during the final 35% of the fourth clock cycle. when the high-side switch is off, the lx node is pulled low by the current flowing through the inductor. this increases the voltage across the bst capacitor. to ensure that the inductor has enough current to pull lx to ground, an internal load sinks current from v out when the device is close to dropout and external load is small. once the input voltage is increased above the dropout region, the device continues to regulate at the set output voltage. the device operates with no load and no external clock at an effective maximum duty cycle of 94% in deep drop - out. this effective maximum duty cycle is influenced by the external load and by the optional external synchro - nized clock. system enable (en) an enable-control input (en) activates the device from the low-power shutdown mode. en is compatible with inputs from the automotive battery level down to 3.3v. the high-voltage compatibility allows en to be con - nected to sup, key/kl30, or the inh inputs of a can transceiver. en turns on the internal regulator. once v bias is above the internal lockout level, v uvl = 3.15v (typ), the control - ler activates and the output voltage ramps up within 2048 cycles of the switching frequency. a logic-low at en shuts down the device. during shut - down, the internal linear regulator and gate drivers turn off. shutdown mode reduces the quiescent current to 9 f a (typ). drive en high to turn on the device.
???????????????????????????????????????????????????????????????? maxim integrated products 11 MAX16975 28v, 1.2a automotive step-down converter with low operating current overvoltage protection the device includes overvoltage protection circuitry that protects the device when there is an overvoltage condi - tion at the output. if the output voltage increases by more than 12% of its set voltage, the device stops switching. the device resumes regulation once the overvoltage condition is removed. overload protection the overload protection circuitry is activated when the device is in current limit and v out is below the reset threshold. under these conditions, the device enters a soft-start mode. when the overcurrent condition is removed before the soft-start mode is over, the device regulates the output voltage to the set value. otherwise, the soft-start cycle repeats until the overcurrent condition is removed. skip mode during light-load operation, i inductor p 200ma, the device enters skip-mode operation. skip mode turns off the internal switch and allows the output to drop below regulation voltage before the switch is turned on again. the lower the load current, the longer it takes for the regulator to initiate a new cycle effectively increasing light-load efficiency. during skip mode, the device qui - escent current drops to as low as 45 f a. overtemperature protection thermal-overload protection limits the total power dis - sipation in the device. when the junction temperature exceeds +175 n c (typ), an internal thermal sensor shuts down the step-down controller, allowing the device to cool. the thermal sensor turns on the device again after the junction temperature cools by +15 n c. applications information output voltage/reset threshold resistive divider network although the devices output voltage and reset threshold can be set individually, figure 1 shows a combined resis - tive divider network to set the desired output voltage and the reset threshold using three resistors. use the follow - ing formula to determine the r fb3 of the resistive divider network: total ref fb3 out r v r v = where v ref = 1v, r total = selected total resistance of r fb1 , r fb2 , and r fb3 in ohms, and v out is the desired output voltage in volts. use the following formula to calculate the value of r fb2 of the resistive divider network: total ref_res fb2 fb3 res r v r r v = ? where v ref_res is 1.25v (see the electrical characteristics table) and v res is the desired reset threshold in volts. the precision of the reset threshold function is depen - dent on the tolerance of the resistors used for the divider. bst capacitor selection for dropout operation the device includes an internal boost capacitor refresh algorithm for dropout operation. this is required to ensure proper boost capacitor voltage that delivers power to the gate-drive circuitry. when the hsfet is on consecutively for 3.65 clock cycles, the internal counter detects this and turns off the hsfet for 0.35 clock cycles. this is of particular concern when v in is falling and approaching v out at the minimum switching frequency (220khz). the worst-case condition for boost capacitor refresh time is with no load on the output. for the boost capacitor to recharge completely, the lx node must be pulled to ground. if there is no current through the inductor then the lx node does not go to ground. to solve this issue, an internal load of about 100ma turns on at the sixth clock cycle, which is determined by a separate counter. in the worst-case condition with no load, the lx node does not go below ground during the first detect of the figure 1. output voltage/reset threshold resistive divider network r fb3 r fb2 r fb1 v out reseti fb MAX16975
???????????????????????????????????????????????????????????????? maxim integrated products 12 MAX16975 28v, 1.2a automotive step-down converter with low operating current 3.65 clock cycles. the device waits for the next 3.65 clock cycles to finish. as a result, the soonest the lx node can go below ground is 4 + 3.65 = 7.65 clock cycles. this time does not factor in the size of the induc - tor and the time it takes for the inductor current to build up to 100ma (internal load). no load minimum time before refresh is: ?t (no load) = 7.65 clock cycles = 7.65 x 4.54s (at 220khz) = 34.73s assuming a full 100ma is needed to refresh the bst capac - itor and depending on the size of the inductor, the time it takes to build up full 100ma in the inductor is given by: ?t (inductor) = l x ?i/?v (current build-up starts from the sixth clock cycle) l = inductor value chosen in the design guide. ?i is the required current = 100ma. ?v = voltage across the inductor (assume this to be 0.5v), which means v in is greater than v out by 0.5v. if ?t (inductor) < 7.65 C 6 (clock cycles) then the bst capacitor is sized as follows: bst_cap i_bst(dropout) x ?t (no load)/?v (bst capacitor) ?t (no load) = 7.65 clock cycles = 34.73s. ?v (bst capacitor), for (3.3v to 5v) output = v out C 2.7v (2.7v is the minimum voltage allowed on the bst capaci - tor). if ?t (inductor) > 7.65 - 6 clock cycles then we need to wait for the next count of 3.65 clock cycles making ?t (no load) = 11.65 clock cycles. assume ?t (no load) to be 16 clock cycles when design - ing the bst capacitor with a typical inductor value for 220khz operation. the final bst_cap equation is: bst_cap = i_bst (dropout) x ?t (no load)/?v (bst capacitor) where: i_bst (dropout) = 2.5ma (worst case) ?t (no load) = 16 clock cycles ?v (bst capacitor) = v out - 2.7v reset timeout period the device offers a capacitor-adjustable reset timeout period. cres can source 10 f a of current. use the fol - lowing formula to set the timeout period. 1.13v c reset_timeout (s) 10 a = where c is the capacitor from cres to gnd in farads. internal oscillator the devices internal oscillator is programmable from 220khz to 1.0mhz using a single resistor at fosc. use the following formula to calculate the switching frequency: 9 osc 26.4 10 ( x hz) f (hz) r ? where r is the resistor from fosc to gnd in ohms. for example, a 220khz switching frequency is set with r fosc = 120k i . higher frequencies allow designs with lower inductor values and less output capacitance. consequently, peak currents and i 2 r losses are lower at higher switching frequencies, but core losses, gate- charge currents, and switching losses increase. inductor selection three key inductor parameters must be specified for operation with the device: inductance value (l), inductor saturation current (i sat ), and dc resistance (r dcr ). to select inductance value, the ratio of inductor peak-to- peak ac current to dc average current (lir) must be selected first. a good compromise between size and loss is a 30% peak-to-peak ripple current to average-current ratio (lir = 0.3). the switching frequency, input voltage, output voltage, and selected lir then determine the inductor value as follows: out supsw out supsw sw out v (v - v ) l v f i lir = where v supsw , v out , and i out are typical values (so that efficiency is optimum for typical conditions). the switching frequency is set by r fosc . the exact inductor value is not critical and can be adjusted to make trade- offs among size, cost, efficiency, and transient response requirements. table 1 shows a comparison between small and large inductor sizes. table 1. inductor size comparison inductor size smaller larger lower price smaller ripple smaller form-factor higher efficiency faster load response larger fixed-frequency range in skip mode
???????????????????????????????????????????????????????????????? maxim integrated products 13 MAX16975 28v, 1.2a automotive step-down converter with low operating current the inductor value must be chosen so that the maximum inductor current does not reach the minimum current limit of the device. the optimum operating point is usually found between 15% and 35% ripple current. when pulse skipping (light loads), the inductor value also determines the load-current value at which pfm/pwm switchover occurs. find a low-loss inductor having the lowest possible dc resistance that fits in the allotted dimensions. most inductor manufacturers provide inductors in standard values, such as 1.0 f h, 1.5 f h, 2.2 f h, 3.3 f h, etc. also look for nonstandard values, which can provide a better compromise in lir across the input voltage range. if using a swinging inductor (where the no-load inductance decreases linearly with increasing current), evaluate the lir with properly scaled inductance values. for the selected inductance value, the actual peak-to-peak inductor ripple current ( d i inductor ) is defined by: out supsw out inductor supsw sw v (v - v ) i v f l ? = where d i inductor is in a, l is in h, and f sw is in hz. ferrite cores are often the best choices, although pow - dered iron is inexpensive and can work well at 220khz. the core must be large enough not to saturate at the peak inductor current (i peak ): inductor peak load(max) i i i 2 ? = + input capacitor the input filter capacitor reduces peak currents drawn from the power source and reduces noise and voltage ripple on the input caused by the circuits switching. the input capacitor rms current requirement (i rms ) is defined by the following equation: out supsw out rms load(max) supsw v (v - v ) i i v = i rms is at a maximum value when the input voltage equals twice the output voltage (v supsw = 2v out ), so i rms(max) = i load(max) /2. choose an input capacitor that exhibits less than +10 n c self-heating temperature rise at the rms input current for optimal long-term reliability. the input-voltage ripple comprises d v q (caused by the capacitor discharge) and d v esr (caused by the esr of the capacitor). use low-esr ceramic capacitors with high ripple-current capability at the input. assume the contribution from d v q and d v esr to be 50%. calculate the input capacitance and esr required for a specified input-voltage ripple using the following equations: esr in l out v esr i i 2 ? = ? + where: supsw out out l supsw sw (v - v ) v i v f l ? = and out in q sw i d(1- d) c v f = ? and out supsw v d v = i out is the maximum output current and d is the duty cycle. output capacitor the output filter capacitor must have low enough equiva - lent series resistance (esr) to meet output ripple and load transient requirements, yet have high enough esr to satisfy stability requirements. the output capacitance must be high enough to absorb the inductor energy while transitioning from full-load to no-load conditions without tripping the overvoltage fault protection. when using high-capacitance, low-esr capacitors, the filter capaci - tors esr dominates the output voltage ripple. so the size of the output capacitor depends on the maximum esr required to meet the output voltage ripple (v ripple(p-p) ) specifications: ripple(p p) load(max) v esr i lir ? = the actual capacitance value required relates to the physical size needed to achieve low esr, as well as to the chemistry of the capacitor technology. thus, the capacitor is usually selected by esr and voltage rating rather than by capacitance value.
???????????????????????????????????????????????????????????????? maxim integrated products 14 MAX16975 28v, 1.2a automotive step-down converter with low operating current when using low-capacity filter capacitors, such as ceramic capacitors, size is usually determined by the capacity needed to prevent v sag and v soar from caus - ing problems during load transients. generally, once enough capacitance is added to meet the overshoot requirement, undershoot at the rising load edge is no longer a problem. however, low-capacity filter capacitors typically have high-esr zeros that can affect the overall stability. rectifier selection the device requires an external schottky diode recti - fier as a freewheeling diode. connect this rectifier close to the device using short leads and short pcb traces. choose a rectifier with a continuous current rating higher than the highest output current-limit threshold (1.5a) and with a voltage rating higher than the maximum expected input voltage, v supsw . use a low forward-voltage-drop schottky rectifier to limit the negative voltage at lx. avoid higher than necessary reverse-voltage schottky rectifiers that have higher forward-voltage drops. compensation network the device uses an internal transconductance error amplifier with its inverting input and its output available for external frequency compensation. the output capacitor and compensation network determine the loop stability. the inductor and the output capacitor are chosen based on performance, size, and cost. additionally, the compen - sation network optimizes the control-loop stability. the controller uses a current-mode control scheme that regulates the output voltage by forcing the required cur - rent through the external inductor, so the device uses the voltage drop across the high-side mosfet. current- mode control eliminates the double pole in the feedback loop caused by the inductor and output capacitor result - ing in a smaller phase shift and requiring less elaborate error-amplifier compensation than voltage-mode control. a simple single series resistor (r c ) and capacitor (c c ) are all that is required to have a stable, high-bandwidth loop in applications where ceramic capacitors are used for output filtering ( figure 2 ). for other types of capaci - tors, due to the higher capacitance and esr, the fre - quency of the zero created by the capacitance and esr is lower than the desired closed-loop crossover fre - quency. to stabilize a nonceramic output capacitor loop, add another compensation capacitor (c f ) from comp to gnd to cancel this esr zero. the basic regulator loop is modeled as a power modula - tor, output feedback divider, and an error amplifier. the power modulator has a dc gain set by g mc o r load , with a pole and zero pair set by r load , the output capacitor (c out ), and its esr. the following equations allow to approximate the value for the gain of the power modulator (gain mod(dc) ), neglecting the effect of the ramp stabilization. ramp stabilization is necessary when the duty cycle is above 50% and is internally done for the device. load sw mod(dc) mc load sw r f l gain g r (f l) = + where r load = v out /i lout(max) in i , f sw is the switch - ing frequency in mhz, l is the output inductance in f h, and g mc = 3s. in a current-mode step-down converter, the output capacitor, its esr, and the load resistance introduce a pole at the following frequency: pmod load sw out load sw 1 f r f l 2 c esr r (f l) = ? ? + ? ? + ? ? the output capacitor and its esr also introduce a zero at: zmod out 1 f 2 esr c = when c out is composed of n identical capacitors in parallel, the resulting c out = n o c out(each) and esr = esr (each) /n. note that the capacitor zero for a paral - lel combination of alike capacitors is the same as for an individual capacitor. figure 2. compensation network r 2 r 1 v ref v out r c c c c f comp g m
???????????????????????????????????????????????????????????????? maxim integrated products 15 MAX16975 28v, 1.2a automotive step-down converter with low operating current the feedback voltage-divider has a gain of gain fb = v fb /v out , where v fb is 1v (typ). the transconductance error amplifier has a dc gain of gain ea(dc) = g m,ea o r out,ea , where g m,ea is the error-amplifier transcon - ductance, which is 1000 f s (typ), and r out,ea is the output resistance of the 50m i error amplifier. a dominant pole (f dpea ) is set by the compensa - tion capacitor (c c ) and the amplifier output resistance (r out,ea ). a zero (f zea ) is set by the compensation resis - tor (r c ) and the compensation capacitor (c c ). there is an optional pole (f pea ) set by c f and r c to cancel the output capacitor esr zero if it occurs near the crossover frequency (f c , where the loop gain equals 1 (0db)). thus: dpea c out,ea c 1 f 2 c (r r ) = + zea c c 1 f 2 c r = pea f c 1 f 2 c r = the loop-gain crossover frequency (f c ) is set below 1/5th the switching frequency and much higher than the power-modulator pole (f pmod ): sw pmod c f f f 5 << the total loop gain as the product of the modulator gain, the feedback voltage-divider gain, and the error-amplifier gain at f c is equal to 1. so: c fb mod(fc) ea(f ) out v gain gain 1 v = for the case where f zmod is greater than f c : ea(fc) m,ea c gain g r = pmod mod(fc) mod(dc) c f gain gain f = therefore: fb mod(fc) m,ea c out v gain g r 1 v = solving for r c : out c m,ea fb mod(fc) v r g v gain = set the error-amplifier compensation zero formed by r c and c c (f zea ) at the f pmod . calculate the value of c c as follows: c pmod c 1 c 2 f r = if f zmod is lower than 5 x f c , add a second capacitor, c f , from comp to gnd and set the compensation pole formed by r c and c f (f pea ) at the f zmod . calculate the value of c f as follows: f zmod c 1 c 2 f r = as the load current decreases, the modulator pole also decreases; however, the modulator gain increases accordingly and the crossover frequency remains the same. for the case where f zmod is less than f c : the power-modulator gain at f c is: c pmod mod(f ) mod(dc) zmod f gain gain f = the error-amplifier gain at f c is: c zmod ea(f ) m,ea c c f gain g r f = therefore: c zmod fb mod(f ) m,ea c out c f v gain g r 1 v f = solving for r c : c out c c m,ea fb mod(f ) zmod v f r g v gain f =
???????????????????????????????????????????????????????????????? maxim integrated products 16 MAX16975 28v, 1.2a automotive step-down converter with low operating current set the error-amplifier compensation zero formed by r c and c c at the f pmod (f zea = f pmod ): c pmod c 1 c 2 f r = if f zmod is less than 5 o f c , add a second capacitor, c f , from comp to gnd. set f pea = f zmod and calculate c f as follows: f zmod c 1 c 2 f r = pcb layout guidelines careful pcb layout is critical to achieve low switching losses and clean, stable operation. use a multilayer board whenever possible for better noise immunity and power dissipation. follow these guidelines for good pcb layout: 1) use a large contiguous copper plane under the device package. ensure that all heat-dissipating com - ponents have adequate cooling. 2) isolate the power components and high-current path from the sensitive analog circuitry. this is essential to prevent any noise coupling into the analog signals. 3) keep the high-current paths short, especially at the ground terminals. this practice is essential for stable, jitter-free operation. make the high-current path com - prising of an input capacitor, high-side fet, inductor, and the output capacitor as short as possible. 4) keep the power traces and load connections short. this practice is essential for high efficiency. use thick copper pcbs (2oz vs. 1oz) to enhance full-load efficiency. 5) route the analog signal lines away from the high- frequency planes. this ensures integrity of sensitive signals feeding back into the device. 6) make the ground connection for the analog and power section close to the device. this keeps the ground current loops to a minimum. in cases where only one ground is used, enough isolation between analog return signals and high power signals must be maintained. ordering information chip information process: bicmos package information for the latest package outline information and land patterns (footprints), go to www.maxim-ic.com/packages . note that a +, #, or - in the package code indicates rohs status only. package drawings may show a different suffix character, but the drawing pertains to the package regardless of rohs status. /v denotes an automotive qualified part. + denotes a lead(pb)-free/rohs-compliant package. *ep = exposed pad. part temp range pin-package MAX16975aaee/v+ -40c to +125c 16 qsop-ep* MAX16975baee/v+ -40c to +125c 16 qsop package type package code outline no. land pattern no. 16 qsop e16+4 21-0055 90-0167 16 qsop-ep e16e+10 21-0112 90-0239
maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a maxim product. no circuit patent licenses are implied. maxim reserves the right to change the circuitry and specifications without notice at any time. maxim integrated products, 120 san gabriel drive, sunnyvale, ca 94086 408-737-7600 17 ? 2011 maxim integrated products maxim is a registered trademark of maxim integrated products, inc. MAX16975 28v, 1.2a automotive step-down converter with low operating current revision history revision number revision date description pages changed 0 3/11 initial release

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